Low Scattering Photodiode-Modulated Probe For Microwave Near-Field Imaging 13
Z
ON
and Z
p
+ Z
OFF
(see Fig.7). In fact, if a simple capacitor model is assumed for the short
dipole in free-space, resonance frequencies of 2.53 GHz and 3.09 GHz can be calculated for the
ON and OFF states respectively. Results from a simulation done with the thin-wire method
of moment are also shown in the figure. In the simulation, the probe is illuminated with a
uniform plane wave in free space. This shows the normalized difference between the squared
scattered field taken 1 cm away from the dipole in the two states. The results also exhibit
a double peak response. In the measurement, the resonance observed in the waveguide
are shifted to lower frequencies. This shift is thought to be due to imperfections in the
construction and uncertainty in the substrate’s constitutive material parameters. Furthermore,
the value of Z
p
in free-space is not the same as in the waveguide where the dipole is interacting
with the metallic walls. Finally, as these reflection differences are obtained by subtracting
very similar measured values, the results are susceptible to measurement and simulation
inaccuracies. Both curves exhibits a maximum sensitivity near the design frequency of 2.45
GHz. Finally, the waveguide measurement process described above was simulated in HFSS.
The reflection coefficient difference shown in Fig. 13 exhibits peaks near 2.7 GHz and 3 GHz. It
should be noted that this curve is derived from differences between S
11
values with a variation
smaller than 5
×10
−5
in magnitude. Therefore, the frequency shift compared to the other two

)at
h =3mm.
8.1 Taking the square root: sign ambiguity removal
When the NF imager operates in monostatic mode, the measured fields are obtained by taking
the square root of the measured data. The square root of a complex signal v
= X
I
+ jX
Q
has two solutions and it is necessary to select the proper one. The procedure might be
straightforward when the measured field takes nonzero values. In this case it is possible to
ensure continuity of the phase distribution in the whole data set. In contrast, sign retrieval is
not an easy task if nulls occur (i.e., E
= 0) at some locations. In these cases, no clear method
has been addressed to choose the sign of the square root correctly. However, a technique was
reported in (Hygate & Nye, 1990) for some particular cases.
In the case of the microstrip line considered here it is well known that transverse electric field
(E
x
in Fig. 14) has a null on the strip’s symmetry plane and a different sign on both sides.
Thus, even if choosing the sign of the electric field on either side is impossible without a priori
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Low Scattering Photodiode-Modulated Probe For Microwave Near-Field Imaging 15
knowledge. It is assumed that when a contour with zero E field is crossed, the phase changes
by π.
9. Probe correction
The short dipole implementing the probe has a finite length. Therefore, the measured data is
not representative of the fields at a point but rather of the integral of the weighted field along
the probe. To take this effect into account, we used the induced e.m.f method for calculating

2|z|
L

u
z
(11)
The measured field is given by the field-current convolution for every point using Equation 12
(see Fig. 16b).
V
probe
= −
1
J
probe
(0)

L
¯
E
i
.u
z
J(z)dl (12)
This equation was used to process the field calculated by HFSS in Fig. 15. The simulations,
after applying convolution, probe correction, are in very good agreement with the
measurements, both in magnitude and in phase plots, which proves the excellent performance
of the probe (see Fig. 15). Within the
±15 mm interval, the average difference between the
simulated (with probe correction) and measured fields was 6.4% in magnitude and 3.2 degrees
in phase. It is worth mentioning that the probe correction does not alter phase information in

, i.e.,
E ∝ S
21
→ E = K
2
S
21
(14)
Using Equations 13 and 14, we can obtain:
E
= K
Δρ
S
21
(15)
The sensitivity of the system to electric field can be given in terms of the minimum possible
reflection coefficient that can be accurately measured, namely Δρ
min
. Consequently, the
sensitivity of the system is simply given by
E
min
= |K
Δρ
min
S
21
| (16)
Spiral inductor
Photodiode

terminated with a matched load. By simulation, we obtained the field incident on the probe
for an incident input power of 1 watt at the DUT’s input port. The same configuration was
then repeated experimentally, that is to say with the probe located at the same point as in the
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Low Scattering Photodiode-Modulated Probe For Microwave Near-Field Imaging 17
simulations. With the probe in this fixed position to keep S
21
constant, the incident power was
reduced with an attenuator until the receiver’s noise floor was reached. The field sensitivity
was then calculated by scaling the E-field value obtained in simulations by the square root of
the threshold power level (in watts) measured experimentally. In the case where the probe was
in the aperture of the horn (large S
21
), the sensitivity was 0.037 V/m. When it was at a height
of 3 mm above the microstrip line (small S
21
), the sensitivity degraded to reach 54.3 V/ m.
This large difference illustrates a weakness of the monostatic configuration for characterizing
non-radiating structures.
11. Optically modulated scatterer (OMS) probes array: Improving measurement
speed in a NF imager
A linear array of seven OMS probes was developed in order to improve the measurement
speed of the NF imager. In the array, the probes are laid in parallel along a line perpendicular
to the probes’ axes (see Fig. 18). The probes were mounted on a piece of planar foam (ε
r
≈1)
with a spacing of 3 cm between the probes. The foam has a thickness of 1.2 cm and is very
rigid. It also prevents the array from vibrating when a very fast measurement is made. The
array is moved mechanically along one direction, while being moved electronically (as well

phase), a current driver exciting and controlling a laser diode, and a digital controller that
generates the reference signal required by an LIA and also that addresses the RF SPDT switch.
This controller also sends commands to the laser diodes modulating the OMS probes. The
whole setup is controlled by a computer software developed using LabView.
12. Validating the NF imager
12.1 Array calibration
It is practically impossible to make a set of identical OMS probes. Differences in the responses
of the probes can be caused by differences in the photodiode characteristics, materials used,
optical fiber/photodiode coupling quality and many other factors (Mostafavi et al., 2005). In
order to quantify these differences in the probes, we performed a simple monostatic field
probing experiment in which the seven probes are set to measure the E field at the same fixed
point near a DUT. The obtained results are then used to compute a complex correction factor
(CF) corresponding to each probe using Equation 17.
CF
=
E
ref
E
Probe#i
; i = 1, 7 (17)
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Low Scattering Photodiode-Modulated Probe For Microwave Near-Field Imaging 19
(a) Near-field imager circuitry (b) The imager receiving part
Fig. 20. (a) Near-field imager microwave circuitry configured for bistatic operation, and (b)
receiving part of the imager incorporating the OMS probe array and a WR-284 rectangular
waveguide.
Probe # CF |CF| ∠CF(de g)
1 0.8704+j0.0218 0.8706 1.4347
2 0.9645-j0.0806 0.9678 4.7724

(i.e., r
i
; i=1, 7). Each probe is also seen by the AA with a different view. Then, the picked-up
signals will not be identical even if all the probes are exposed to the same fields. So, we
need to compensate the measured data (raw data) for the NF radiation pattern of the AA.
In principle, the simple compensation method described in the previous subsection should
suffice. In practice however it has been observed that the coupling between each probe and the
AA slightly varies when the probes are moved near the AUT, even if the AA is maintained at
a fixed position with respect to the probe array. This variation comes from mutual interaction
of the AA and probe array with the AUT, which is not constant during the scan. A method to
compensate for this effect is introduced in the next paragraph. We first set the AA in Fig. 20b
(a) Magnitude (b) Phase
Fig. 21. The measurement result obtained in the test to compensate for the radiation pattern
of the receiving antenna; (a) magnitude and (b) phase of the normalized measured E field by
the AA in the monostatic setup.
to operate as an illuminator in a monostatic mode (TX/RX device). During this test the AUT
is passive and terminated with a matched load. In this experiment, the probes are addressed
successively and then moved to a new position until the array scans the region of interest
above the AUT. Ideally, a flat response is expected over the region scanned by each probe, but
given the interaction of the array with the surrounding objects, including the passive AUT,
and the interaction between probes, the measured results are not constant, as illustrated in
Fig. 21. In this test the AUT was a horn antenna and the array was scanned at a height of
30 mm (i.e., λ/4) above the aperture. The ideal results, i.e., with no interaction between the
probes with the AUT and the AA are shown by broken line in Fig. 21. The asymmetry of the
curves occurs because of discrepancies in the probes of the array, displacement of probes and
misalignment. Even though each probe is at a fixed distance and angle from the receiving
antenna, significant variation can be observed when a 30 mm interval is scanned. The results
also demonstrate the importance of the compensation before any comparison is made to
validate the imager’s results. After this test, the E-field measurements of the AA at each
98

return-loss of about 12 dB; the physical dimensions of the PIFA are as follows: L
p
=27 mm,
W
p
=13 mm, H
p
=7 mm, P
exc
=7 mm, W
GND
=70 mm and L
GND
=137 mm.
In all cases the measured phase information in the E-plane of the PIFA are in good agreement
over the whole x interval. In order to quantify the difference between the measurement results
and the simulated distribution of the PIFA, the mean square error of the data was calculated.
The error associated with E-plane and H-plane cuts are 0.12% and 0.06%, respectively, with
respect to simulations. The benefit of probe correction in the bistatic case is clearly visible in
the H-plane results.
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Low Scattering Photodiode-Modulated Probe for Microwave Near-Field Imaging
22 Photodioes
(a) Magnitude (b) Phase
Fig. 23. 2-D map of electric field distribution measured (compensated results) at a distance of
λ/4 above AUT; (a) magnitude (dB) and (b) phase (deg.).
(a) E-plane (magnitude) (b) E-plane (phase)
(c) H-plane (magnitude) (d) H-plane (phase)
Fig. 24. E- and H-plane cuts of the measured E-field at distance of λ/4 from the PIFA
antenna’s ground plane; (a) and (c) magnitude (dB), and (b) and (d) phase (deg.).

This chapter addressed and discussed the design and implementation of a NF imager
based on the modulated scatterer technique (MST) in detail. The imager consists of several
optically modulated scatterer (OMS) probes that are very accurate, highly sensitive and also
frequency selective. Each OMS probe was optimized to operate at 2.45 GHz. This probe,
guarantees almost perturbation-free measurements. It can be implemented with low-cost
commercial off-the-shelf photodiodes. The OMS probes were also studied and verified
for omnidirectionality and cross-polarization rejection performance. The probes showed
an absolute deviation of about
±0.3 dB with respect to an omnidirectional response. The
co-to-cross polarization ratio was measured and found to be better than 60 dB. The frequency
response of the probe was studied theoretically and experimentally in order to qualify the
performance of the matching network and to assess its impact on the frequency response of the
OMS probes. The performance of the probes was validated by measuring the NF distribution
of a 50 Ω microstrip transmission line. The measurements were compared with results of
simulations using HFSS. The results also showed that the sensitivity of the measurement
system can be as low as 0.037 V/m. Error associated with magnitude and phase measurement
of respectively 6.4% and 3.2 degrees were observed. By developing a linear array of OMS
probes, the measurement speed for an E-field measurement was increased more than 14
times compared to that obtained with a commercially available opto-mechanically switched
systems. To improve the accuracy of measurements using the array, the raw measurement
data were corrected using the proposed calibration technique, to compensate for uncertainties
in the probes’ responses. The E-field measurements made with the developed imager were
in good agreement with the simulations and were very rapid. Benefiting from carrier
101
Low Scattering Photodiode-Modulated Probe for Microwave Near-Field Imaging
cancellation, the isolation between the input and output ports of the imager was improved
by about 60 dB. This enabled us to increase the signal fed to the NF imager and reach an
overall dynamic range of 85 dB, i.e., an increase of 18 dB compared to the case when the NF
imager is not equipped with the carrier cancellation circuit.
16. References

Iigusa, K., Sawaya, T., Taromaru, M., Ohira, T. & Komiyama, B. (2006). Experimental
proof of electrically invisible state of inductively loaded dipole and proposal of
electrically invisible meander-lines, Antennas and Propagation, IEEE Transactions on
54(11): 3374–3382.
Justice, R. & Rumsey, V. (1955). Measurement of electric field distributions, Antennas and
Propagation, IRE Transactions on 3(4): 177–180.
King, R. J. (1978). Microwave Homodyne System, Wiley New York.
Laurin, J J., Zurcher, J F. & Gardiol, F. (2001). Near-field diagnostics of small printed antennas
using the equivalent magnetic current approach, Antennas and Propagation, IEEE
Transactions on 49(5): 814–828.
Liang, W., Hygate, G., Nye, J., Gentle, D. & Cook, R. (1997). A probe for making
near-field measurements with minimal disturbance: the optically modulated
scatterer, Antennas and Propagation, IEEE Transactions on 45(5): 772–780.
Mostafavi, M., Bolomey, J C. & Picard, D. (2005). Experimental study on compensation
of array element pattern of collinear dipole array sensor, EICE Trans Commun, Inst
Electron Inf Commun Eng. E88-B(8): 3314–3316.
Munoz, K., Perrey, A. & Zoughi, R. (2008). Potential application of the modulated
scatterer technique to multilayered material evaluation and health monitoring,
Instrumentation and Measurement Technology Conference Proceedings, 2008. IMTC 2008.
IEEE, pp. 1643–1647.
Nye, J. (2003). A simple method of spherical near-field scanning to measure the far fields
of antennas or passive scatterers, Antennas and Propagation, IEEE Transactions on
51(8): 2091–2098.
Omrane, B., Laurin, J J. & Goussard, Y. (2006). Subwavelength-resolution microwave
tomography using wire grid models and enhanced regularization techniques,
Microwave Theory and Techniques, IEEE Transactions on 54(4): 1438–1450.
Petre, P. & Sarkar, T. (1992). Planar near-field to far-field transformation using an
equivalent magnetic current approach, Antennas and Propagation, IEEE Transactions
on 40(11): 1348 –1356.
Quilez, M., Aragon, M., Atienza, A., Fernandez-Chimeno, M., Riu, P. & Silva, F.

role in many applications, such as microsurgery, micromachining, laser ranging, nonlinear
optics etc. Methods enabling precise measurement of such pulses duration are therefore
essential. Common silicon photodiodes used in combination with ordinary oscilloscopes
cannot be used for such a precise measurement because of low temporal resolution of the
whole system. Therefore, several sophisticated methods based on optical or electro-optical
effects were developed in the past decades and are still widely used (Diels, 1995; Keller,
2007; Rulliere, 2003). The first method based on nonlinear optical effect is measurement of
second order autocorrelation function of the measured light pulse. This function is always
symmetric and does allow to obtain detailed information on the exact pulse shape and pulse
duration is calculated assuming the certain pulse temporal profile. This method, which can
be used either for single or repetitive pulses, is in principle very precise but has also several
disadvantages. The whole measuring system has to be aligned precisely and the measured
beam has to enter the system in accurately aligned angle. Precision depends also on control
of the delay line used for scanning autocorrelation measurement. Because the resulting pulse
length is in this case calculated from many points, it is clear that it represents average value of
the real pulses lengths and is difficult to evaluate pulse duration stability or study some other
effects, i.e. pulse shortening under specific conditions. The single shot autocorrelators on
the other hand can measure the autocorrelation function from only single laser shot but exact
interpretation is not also unique. The only direct picosecond pulses measurement method
based on electro optical effect uses a streak camera. This method allows to measure single
pulse duration and shape but several consecutive pulses in the pulse train cannot be measured
with sufficient temporal resolution. It is also necessary to carefully align whole measuring
system and perform its rigorous calibration including readout.
Repetitive signals as mode locked pulse trains from continuously pumped lasers can be
measured using sampling oscilloscopes in combination with fast detectors enabling resolution
in units of picoseconds. For signals with repetition rates below several kHz sampling
oscilloscopes cannot be used and only real time oscilloscopes with lower resolution are
available. In the past few years fast real time oscilloscopes with analog bandwidth up to
6
2 Will-be-set-by-IN-TECH

corresponding impulse response, but moreover to investigate pulse duration stability using
oscilloscope sophisticated statistics functions. In addition, two tested laser systems were
operated under special regime resulting in pulse duration shortening along the output pulse
train, which was possible to study precisely at each individual laser shot.
2. Physics of detection and photodiodes
Photodiodes represent one of fundamental light detection devices and play almost
un-substitutable role in many applications, where the time and amplitude characteristic of
the incoming light pulses has to be investigated or further exploited. Among the main
advantages of photodiodes belong ease of use, fast time response, sensitivity at sufficiently
wide spectral region, reasonably low thermal and electrical noise, and small dimensions
enabling integration in electro-optics devices. Photodiodes are mostly used for detection of
laser light pulses in many applications, such as telecommunications, sensing, security, and
laser systems monitoring.
Light detection by a semiconductor material is based on the well-known phenomena of
photon absorption (Saleh, 2007). If the incident photon energy exceeds the band gap energy
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Single Shot Diagnostics of Quasi-Continuously Pumped Picosecond Lasers Using Fast Photodiode and Digital Oscilloscope 3
of the semiconductor, an electron-hole pair is released. These charge carriers move by
diffusion to places with lower concentration and may contribute to electrical current, if the
semiconductor is connected into some electrical circuit. Since the diffusion rate is slow, these
photo-conductors cannot be used as sufficiently fast photodetectors.
The situation changes when a photodiode formed by PN junction is used. Between the P
and N regions, there is a depletion layer consisted of positively and negatively charged fixed
ions. These ions form electrical field in the direction from N to P. If the photon is absorbed
in this depletion layer, the electrons and holes are accelerated by the described electrical field.
In this layer the carriers move by drift process which depends on the electrical field and is
much quicker than diffusion. The depletion layer can be further extended applying reverse
bias voltage on the photodiode which significantly decreases the carrier transit time. Reverse
bias voltage (below the breakdown threshold) increases the noise component formed by dark

photodiodes can be used for wavelengths only up to
∼1.1 μm. There is also second
well-known semiconductor-compound material - gallium arsenide (GaAs). In the pure form,
its long-wavelength region is limited to about 850 nm. By adding other component, the
wavelength region can be significantly extended. Usage of many compounds has been
published but in the commercially available photodiodes mainly Indium (In) component is
used in the discussed spectral region. Pure InAs has λ
cutof f
of about 3.4 μm and depending
on its concentration in In
x
Ga
1−x
As compound, the wavelength range can be tuned. Mainly
used compound has x
∼0.5 determining λ
cutof f
of ∼1.7 μm (Bitter, 2000).
Usage of GaAs-based material for the PIN photodiode construction has also other advantage
in the charge carrier mobility. The electron mobility in GaAs is about five times higher
than in Si while the hole mobility is comparable (Gibbons, 1987). As for construction
parameters, it has already been said that in order to obtain fast response time it is necessary
to keep the depletion layer capacitance as low as possible. Because of the mentioned
reasons commercially available photodiodes based on GaAs / InGaAs have higher frequency
107
Single Shot Diagnostics of Quasi-Continuously
Pumped Picosecond Lasers Using Fast Photodiode and Digital Oscilloscope
4 Will-be-set-by-IN-TECH
bandwidth in comparison with the silicon photodiodes. The fastest photodiodes for
telecommunication have to be based on InGaAs because of its desirable spectral response

simultaneously due to the decoupling of the efficiency from the absorption layer thickness.
Efficient operation of these photodiodes based on the thin film germanium on silicon was
successfully demonstrated (Wang, 2008).
Another approach leading to high-speed photodetectors is based on the metal-semiconductor
Schottky junction, mainly developed in the MSM (Metal-Semiconductor-Metal) photodiode
(Berger, 1996; Kache, 2005). Its structure is comprised of back-to-back Schottky diodes that
use an interdigitated electrode configuration on the top of the active absorbing layer. This
construction leads to low capacitance in comparison with the standard PIN photodiodes and
therefore the MSM photodiode response speed is mostly limited by the transit time of the
photo-generated carriers. Different materials may serve as the active layer and besides IR and
visible spectral range also UV detectors were demonstrated successfully (Liu, 2010).
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Single Shot Diagnostics of Quasi-Continuously Pumped Picosecond Lasers Using Fast Photodiode and Digital Oscilloscope 5
3. Picosecond pulse measurement using oscilloscope - photodiode system
3.1 Measuring system response
This chapter is aimed at measurement of ultrashort light pulses with duration between 10 and
200 ps generated by mode-locked lasers. The measurement is performed using oscilloscope
- photodiode system and therefore overall response time of these both components has to
be treated. Response time of each electrical component is limited by its electrical frequency
bandwidth which is usually defined at the frequency where a sinusoidal output signal
amplitude is attenuated to about 70 % of its original value (or in other terms the signal power
is attenuated to 50 %), also known as -3 dB point. Let’s consider the pulse width (duration) as
full width at half maxima (FWHM) of light intensity. For the pulse width measurement the
laser pulse shape can be in good approximation considered as Gaussian and therefore sum of
square calculation of the real pulse duration FWHM
REA L
can be used:
FWHM
REA L

response) and FWHM
PD
is the photodiode minimal pulse width. Into this calculation also
influence of cables and connectors can be included when their frequency bandwidth cannot
be neglected in comparison with other measuring components bandwidth. Similar theorem
can be used for the calculation of rise time (RT) as the system response on the step input signal
(Johnson, 1993; Keller, 2007).
3.2 Minimal pulse width and rise time of the measuring system components
Minimal pulse width FWHM and rise time RT of the measuring components can be calculated
for the given frequency bandwidth f
3dB
according to the formulas
FWHM
=
K
FWHM
f
3dB
, RT =
K
RT
f
3dB
(3)
Constant K varies in range from 0.3 to 0.5 according to the step or impulse response and also
according to calculation performed for the oscilloscope or photodiode.
3.2.1 Photodiode response
The photodiode rise time can be calculated for the given electrical bandwidth using K
RT
= 0.35.

K
RT
its value can be just estimated.
3.2.3 Calculation of the measuring system impulse response
According to the previous analysis, the step and impulse response of the measuring system
used in our experiments can be calculated. The measuring system consisted of the PIN
photodiode EOT ET-3500 (EOT, 2011) connected by high frequency SMA cable to the real
time oscilloscope LeCroy SDA-9000 (LeCroy, 2009)
1
. Datasheet parameters of the photodiode
are as following: cutoff frequency >15 GHz, rise and fall time <25 ps, spectral range 1000 -
1650 nm, responsivity 0.88 A/W @ 1550 nm, active area diameter 32 μm, junction capacitance
0.12 pF. Important datasheet parameters as well as rise time and minimal pulse width
calculation are summarized in Table 1.
Compo- Datasheet values Step response Impulse response
nents & f
3dB
RT RT K
FWHM
FWHM K
FWHM
FWHM
System [GHz] [ps] [ps] [ps] [ps]
LeCroy 9.0 <49 49 0.44 49 0.40 45
EOT 15.0 <25 25 0.31 21 0.31 21
System - - 55 - 53 - 50
Table 1. Datasheet values and calculated system response FWHM
SY STEM
.
The system rise time can be calculated directly from the given datasheet values. The minimal